The basic principle of using time-of-flight (TOF) measurements for range finding applications is to measure how long it takes for radiation, e.g. photons to travel over an unknown distance. The unknown distance can then be deduced from the measured time of flight in combination with the known speed of the radiation such as light.
Many ways of how to modulate a light source for such TOF measurements, and which strategy to follow for making the distance measurement are known to a person skilled in the art and are described in patents and scientific literature. Most of these range-finding systems use a receiver in which a mixer is used to demodulate an incoming photocurrent for finding e.g. a phase or a time period for distance estimation. The photocurrent is typically mixed with a reference signal.
A problem is to separate signals originating from background light efficiently from signals originating from useful TOF-light. The useful TOF-light may be emanating from a modulated light source. The background light that is present on an area in a scene, of which the distance is to be measured, can be a plurality of orders of magnitude larger, e.g. up to six orders of magnitude larger, than the light present on this same area and originating from the modulated light source. It is known from literature how to reduce this large difference to some extent by using an optical filter, which attenuates the visible background light from the TOF light based on wavelength differences. In this way a reduction of an order of magnitude can be obtained. With a narrow-band optical pass filter and using a narrow-band laser light source for generating the useful TOF-light, possibly two orders of magnitude can be overcome. However, LED light sources are preferred light sources for future TOF range finders, since there brightness may be much higher; they may emit Watts of light, whereas lasers may only emit milli-Watts of light in free space for eye-safety reasons.
In U.S. Pat. No. 7,268,858 a device is described to measure TOF signals in the presence of large background light signals. This is done using transistors to continuously compensate for the extra drawn background light current. Therefore this solution adds transistor noise to the circuit, thus degrading the signal to noise ratio and distance accuracy.
A better signal to noise ratio is obtained by accumulating the charges on a capacitance for a specified time interval, then sampling the capacitor value and resetting it for a subsequent measurement. This is a well known technique used in standard imaging, e.g. in active pixels; it will be referred to hereinafter as capacitance integration. Major drawback of the approach is the limited upper end dynamic range due to saturation of the capacitor. Capacitance integration has been reused in TOF imaging to similarly optimize signal to noise ratio, but unfortunately, due to this limited upper end dynamic range for both useful and background light, rapidly fails in the presence of background light. This is illustrated in FIG. 2A and FIG. 2B. FIG. 2A shows exemplary transient output signals of a typical prior art time-of-flight sensor, implementing capacitance integration. In these sensors mixing is typically carried out asymmetrically by multiplying to 1 and 0, not to 1 and −1. Therefore background light translates to a signal that contributes equally to both output signals 10 and 11. The TOF useful signal is contained in the difference 12. When, as shown in FIG. 2B, a larger background light component is present, the output signals, 14 and 15, saturate to ground before being sampled at tsample and lose the needed difference information. A straight forward way to solve this is shortening the integration interval or sampling the signal earlier, for example at tsample2. This, however, results in a smaller measured difference amplitude 13 and in a smaller signal-to-noise ratio.
For use in standard imaging, techniques have been developed to stretch the limited upper end dynamic range of these capacitance integrating circuits. In FIG. 1A a general schematic of such a circuit is shown, as discussed in e.g. U.S. Pat. No. 6,130,713. Inside each pixel an automatic reset circuit 100 and a counter 4 are provided. A comparator 3 of the automatic reset circuit 100 triggers reset transistor 1 whenever a predefined threshold value Vref is passed, resetting detector node 2 and shifting the voltage over reset voltage 5 (see FIG. 1B) so that integration is restarted. The counter 4 counts the total number of resets N taking place during one integration period. In FIG. 1B the voltage evolution 6 on detector node 2 is shown. At the end of the integration time, at moment tsample, output value 8 is obtained together with the number N of resets. The total intensity value is found by multiplying counter output N with reset voltage 5 and adding the sampled output value 8. If the added reset noise (noise generated by the reset transistor 1) is ignored the dynamic range is in this way extended N times.
This technique, implementing capacitance integration with a dynamic range extension circuit has also been reused in time of flight detector read-out circuits, for example in U.S. Pat. No. 6,919,549 and U.S. Pat. No. 7,157,685, resulting in the same extension of the dynamic range and thus, up to some degree, in an increased tolerance to background light. However, since in these techniques the amount of needed circuitry scales with the achieved dynamic range extension, for obtaining reasonable background light suppression a very large substrate area, e.g. silicon area, is needed.
Other dedicated, but more complex, time-of-flight techniques extending dynamic range and thus tolerating background light up to some degree exist, like for example the techniques claimed in U.S. Pat. No. 7,176,438 or in U.S. Pat. No. 6,678,039.
Therefore, though many techniques exist in the field, there is still room for improvement.